I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature
of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here... >http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but too
far and you have oscillation. Also, problems occur with slew rate
limiting due to Cdom, the TR5 constant current, and the increase in hFE
of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead. Seems
stable and not slew rate limiting. Took a lot longer to do than that >sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8 probably
get hotter and their Vbe would drop by more than the diodes, so it's
class B. As it runs at a high temperature, I obviously want to reduce >dissipation, and if I'd used say a rubber diode to get some quiescent >current, I think it would be very difficult to control Iq well enough
over the temperature range.
So I have a circuit which works well enough, but could be better with
regard to crossover distortion (though it's lower than I would have
thought). In this application, the better the signal quality, the
higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature
of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here... >http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but too
far and you have oscillation. Also, problems occur with slew rate
limiting due to Cdom, the TR5 constant current, and the increase in hFE
of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead. Seems
stable and not slew rate limiting. Took a lot longer to do than that >sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8 probably
get hotter and their Vbe would drop by more than the diodes, so it's
class B. As it runs at a high temperature, I obviously want to reduce >dissipation, and if I'd used say a rubber diode to get some quiescent >current, I think it would be very difficult to control Iq well enough
over the temperature range.
So I have a circuit which works well enough, but could be better with
regard to crossover distortion (though it's lower than I would have
thought). In this application, the better the signal quality, the
higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
On a sunny day (Wed, 6 Dec 2023 15:26:00 +0000) it happened Clive Arthur <clive@nowaytoday.co.uk> wrote in <ukq3qb$qjik$1@dont-email.me>:<snip>
I'm not an analog design expert, but needs must.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
My experience with this sort of transistor amplifiers is that it depends a lot on what transistors and what manufacturer
you use.
Hard to tell this way
In the 3055 days one make transistor oscillated, the other was OK.
The temperatiure range you mention is extreme, not much power left at 180C! Huge heatsink?
How much power output do you need?
On Wed, 6 Dec 2023 15:26:00 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature
of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here...
http://www.douglas-self.com/ampins/dipa/dipa.htm
That's really ancient. And what is fig 1b all about?
What are your requirements? Current, distortion, protections? What's
your load?
It won't work if you call the transistors TR; they feel insulted and oscillate. They want to be called Q.
On Wednesday, December 6, 2023 at 11:45:32?AM UTC-5, Clive Arthur wrote:
On 06/12/2023 16:19, John Larkin wrote:
On Wed, 6 Dec 2023 15:26:00 +0000, Clive ArthurIt's old, but discrete and can take some pain. Too many IC's have
<cl...@nowaytoday.co.uk> wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature >> >> of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here...
http://www.douglas-self.com/ampins/dipa/dipa.htm
That's really ancient. And what is fig 1b all about?
thermal limiting. Fig 1b? dunno, I just Googled the circuit diagram as
an example of the architecture.
Fig 1b is Class A emitter follower,
Fig 1a is a Class B, also emitter follower pair.
The TR8/9 and TR6/7 are well known composite emitter follower configurations. They should be wideband and not significantly affect loop phase at 100KHz. Dunno how you get crossover distortion with 1A, unless your 'VBIAS' is be too slight.
What exactly does '100KHz" refer to? Is this a bandwidth or CW frequency?
Do you have that right about 180oC? Seems extreme.
What are your requirements? Current, distortion, protections? What'sIt's a few watts, coupled capacitively to various very lossy lines,
your load?
maybe 30R. I just want to know if there are any bright ideas for
improving it within the constraints.
It won't work if you call the transistors TR; they feel insulted andNo, PNP's are pink and NPN's are blue. I'll have none of this LGBTQ
oscillate. They want to be called Q.
nonsense.
<snip>
--
Cheers
Clive
On 06/12/2023 16:19, John Larkin wrote:
On Wed, 6 Dec 2023 15:26:00 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature >>> of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here...
http://www.douglas-self.com/ampins/dipa/dipa.htm
That's really ancient. And what is fig 1b all about?
It's old, but discrete and can take some pain. Too many IC's have
thermal limiting. Fig 1b? dunno, I just Googled the circuit diagram as
an example of the architecture.
What are your requirements? Current, distortion, protections? What's
your load?
It's a few watts, coupled capacitively to various very lossy lines,
maybe 30R. I just want to know if there are any bright ideas for
improving it within the constraints.
It won't work if you call the transistors TR; they feel insulted and
oscillate. They want to be called Q.
No, PNP's are pink and NPN's are blue. I'll have none of this LGBTQ >nonsense.
<snip>
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature
of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here... http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but too
far and you have oscillation. Also, problems occur with slew rate
limiting due to Cdom, the TR5 constant current, and the increase in hFE
of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead. Seems stable and not slew rate limiting. Took a lot longer to do than that sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8 probably
get hotter and their Vbe would drop by more than the diodes, so it's
class B. As it runs at a high temperature, I obviously want to reduce dissipation, and if I'd used say a rubber diode to get some quiescent current, I think it would be very difficult to control Iq well enough
over the temperature range.
So I have a circuit which works well enough, but could be better with
regard to crossover distortion (though it's lower than I would have thought). In this application, the better the signal quality, the
higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
On 06/12/2023 15:26, Clive Arthur wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature
of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here...
http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but too
far and you have oscillation. Also, problems occur with slew rate
limiting due to Cdom, the TR5 constant current, and the increase in hFE
of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead. Seems
stable and not slew rate limiting. Took a lot longer to do than that
sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8 probably
get hotter and their Vbe would drop by more than the diodes, so it's
class B. As it runs at a high temperature, I obviously want to reduce
dissipation, and if I'd used say a rubber diode to get some quiescent
current, I think it would be very difficult to control Iq well enough
over the temperature range.
So I have a circuit which works well enough, but could be better with
regard to crossover distortion (though it's lower than I would have
thought). In this application, the better the signal quality, the
higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
Maintaining class AB bias over that temperature range is going to be >difficult. I'd look instead at having no Vbias so the power stage
operates pure class B and then have a fast small class A stage fill in
the cross-over distortion. In other words the Quad feed-forward aka
current dumping idea of the 1970s.
piglet
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature
of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here... http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but too
far and you have oscillation. Also, problems occur with slew rate
limiting due to Cdom, the TR5 constant current, and the increase in hFE
of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead. Seems stable and not slew rate limiting. Took a lot longer to do than that sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8 probably
get hotter and their Vbe would drop by more than the diodes, so it's
class B. As it runs at a high temperature, I obviously want to reduce dissipation, and if I'd used say a rubber diode to get some quiescent current, I think it would be very difficult to control Iq well enough
over the temperature range.
So I have a circuit which works well enough, but could be better with
regard to crossover distortion (though it's lower than I would have thought). In this application, the better the signal quality, the
higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
On 2023-12-06 10:26, Clive Arthur wrote:
I'm not an analog design expert, but needs must.Some local feedback around the output stage would get my vote. The
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient
temperature of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here...
http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but too
far and you have oscillation. Also, problems occur with slew rate
limiting due to Cdom, the TR5 constant current, and the increase in
hFE of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead. Seems
stable and not slew rate limiting. Took a lot longer to do than that
sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8 probably
get hotter and their Vbe would drop by more than the diodes, so it's
class B. As it runs at a high temperature, I obviously want to reduce
dissipation, and if I'd used say a rubber diode to get some quiescent
current, I think it would be very difficult to control Iq well enough
over the temperature range.
So I have a circuit which works well enough, but could be better with
regard to crossover distortion (though it's lower than I would have
thought). In this application, the better the signal quality, the
higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
Sziklai pairs have their own local feedback, but that doesn't fix the crossover problem.
Another approach would be to turn TR4 into a diff pair. TR3's collector swing is going to waste, and that would let you keep the open-loop gain
the same, while stabilizing the tail current.
On 2023-12-06 10:26, Clive Arthur wrote:
I'm not an analog design expert, but needs must.Some local feedback around the output stage would get my vote. The
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient
temperature of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here...
http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but too
far and you have oscillation. Also, problems occur with slew rate
limiting due to Cdom, the TR5 constant current, and the increase in
hFE of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead. Seems
stable and not slew rate limiting. Took a lot longer to do than that
sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8 probably
get hotter and their Vbe would drop by more than the diodes, so it's
class B. As it runs at a high temperature, I obviously want to reduce
dissipation, and if I'd used say a rubber diode to get some quiescent
current, I think it would be very difficult to control Iq well enough
over the temperature range.
So I have a circuit which works well enough, but could be better with
regard to crossover distortion (though it's lower than I would have
thought). In this application, the better the signal quality, the
higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
Sziklai pairs have their own local feedback, but that doesn't fix the crossover problem.
Another approach would be to turn TR4 into a diff pair. TR3's collector swing is going to waste, and that would let you keep the open-loop gain
the same, while stabilizing the tail current.
Cheers
Phil Hobbs
To reduce the tempco of the first stage's tail source, one approach
would be to use a LED or a SiC diode instead of two silicon ones in series.
I've sometimes used a LED with an emitter follower to make a temperature-compendated low noise 1-V reference. You have to futz
around with choosing LEDs to make that work well, but some orange Avago
LEDs got down to a couple of hundred microvolts/K.
I don't know whether there are LEDs that survive long enough in your conditions. Alternatively, the GB01SLT06-214 SiC rectifier seems to
have about the same -2 mV dV/dT as a BJT, while dropping more voltage,
so that might work.
Cheers
Phil Hobbs
(Who still wants to make a thermoacoustic fridge to make downhole instruments' lives easier.)
On 06/12/2023 21:19, Phil Hobbs wrote:
On 2023-12-06 10:26, Clive Arthur wrote:
I'm not an analog design expert, but needs must.Some local feedback around the output stage would get my vote. The
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient
temperature of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here...
http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but too
far and you have oscillation. Also, problems occur with slew rate
limiting due to Cdom, the TR5 constant current, and the increase in
hFE of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead. Seems
stable and not slew rate limiting. Took a lot longer to do than that
sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8 probably
get hotter and their Vbe would drop by more than the diodes, so it's
class B. As it runs at a high temperature, I obviously want to reduce
dissipation, and if I'd used say a rubber diode to get some quiescent
current, I think it would be very difficult to control Iq well enough
over the temperature range.
So I have a circuit which works well enough, but could be better with
regard to crossover distortion (though it's lower than I would have
thought). In this application, the better the signal quality, the
higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
Sziklai pairs have their own local feedback, but that doesn't fix the
crossover problem.
Another approach would be to turn TR4 into a diff pair. TR3's
collector swing is going to waste, and that would let you keep the
open-loop gain the same, while stabilizing the tail current.
Cheers
Phil Hobbs
Thanks, that second one particularly sounds like a good idea. I'll see
what the sim says. Local feedback on the output stage sounds trickier,
I'll have to think.
One thing I thought of was to use multiple smaller output pairs in
parallel, and have a DC offset for each one. Imagine replacing the
Vbias diode with a string of a few series diodes, and connecting the
bases of one output pair across D1, the next pair across D2 etc. Would
need resistors from each pair of emitters to the output. That should
give lower crossover in more places. Maybe.
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature
of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here... >http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but too
far and you have oscillation. Also, problems occur with slew rate
limiting due to Cdom, the TR5 constant current, and the increase in hFE
of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead. Seems
stable and not slew rate limiting. Took a lot longer to do than that >sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8 probably
get hotter and their Vbe would drop by more than the diodes, so it's
class B. As it runs at a high temperature, I obviously want to reduce >dissipation, and if I'd used say a rubber diode to get some quiescent >current, I think it would be very difficult to control Iq well enough
over the temperature range.
So I have a circuit which works well enough, but could be better with
regard to crossover distortion (though it's lower than I would have
thought). In this application, the better the signal quality, the
higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
Some local feedback around the output stage would get my vote. The
Sziklai pairs have their own local feedback, but that doesn't fix the >crossover problem.
On 06/12/2023 15:59, Jan Panteltje wrote:
On a sunny day (Wed, 6 Dec 2023 15:26:00 +0000) it happened Clive Arthur<snip>
<clive@nowaytoday.co.uk> wrote in <ukq3qb$qjik$1@dont-email.me>:
I'm not an analog design expert, but needs must.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
My experience with this sort of transistor amplifiers is that it depends a lot on what transistors and what manufacturer
you use.
Hard to tell this way
In the 3055 days one make transistor oscillated, the other was OK.
The temperatiure range you mention is extreme, not much power left at 180C! >> Huge heatsink?
How much power output do you need?
A few watts, the heatsink isn't huge, there's no room. Tj on the output
pair is probably around 210'C at 180'C ambient. It doesn't come with a >lifetime guarantee.
Would it not be simpler to use the signal to FM modulate say a 100 MHz carrier
and detect that at the other end?
On 07/12/2023 08:57, Jan Panteltje wrote:
<snip>
Would it not be simpler to use the signal to FM modulate say a 100 MHz carrier
and detect that at the other end?
The 'cable' is horrible. There are many different types, absolutely and >completely non-negotiable, but for the longest, getting 100kHz through
is often not achievable. The signal looks like white noise. Many
sinusoids from 5kHz to 100kHz.
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature
of -20'C to 180'C.
On Wed, 6 Dec 2023 15:26:00 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature
of -20'C to 180'C.
No you haven't. Not using conventional components.
RL
On 06/12/2023 21:19, Phil Hobbs wrote:
On 2023-12-06 10:26, Clive Arthur wrote:
I'm not an analog design expert, but needs must.Some local feedback around the output stage would get my vote. The
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient
temperature of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here...
http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but too
far and you have oscillation. Also, problems occur with slew rate
limiting due to Cdom, the TR5 constant current, and the increase in
hFE of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead. Seems
stable and not slew rate limiting. Took a lot longer to do than that
sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8 probably
get hotter and their Vbe would drop by more than the diodes, so it's
class B. As it runs at a high temperature, I obviously want to reduce
dissipation, and if I'd used say a rubber diode to get some quiescent
current, I think it would be very difficult to control Iq well enough
over the temperature range.
So I have a circuit which works well enough, but could be better with
regard to crossover distortion (though it's lower than I would have
thought). In this application, the better the signal quality, the
higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
Sziklai pairs have their own local feedback, but that doesn't fix the
crossover problem.
On Wed, 6 Dec 2023 15:26:00 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V >>single supply, and to go up to 100kHz with a working ambient temperature
of -20'C to 180'C.
No you haven't. Not using conventional components.
RL
On 07/12/2023 15:04, legg wrote:
On Wed, 6 Dec 2023 15:26:00 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature >>> of -20'C to 180'C.
No you haven't. Not using conventional components.
RL
Bugger! I could have sworn it was working at 180'C (along with all the
other parts of the system), but it seems you know better. I must have a >faulty oven.
I'd better warn all the other downhole instrumentation companies too!
But yes, selected conventional components, analog and digital. And yes,
I know that if you extrapolate the graphs, most of the parts de-rate to >negative power dissipation.
On Thu, 7 Dec 2023 15:26:52 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
On 07/12/2023 15:04, legg wrote:
On Wed, 6 Dec 2023 15:26:00 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V >>>> single supply, and to go up to 100kHz with a working ambient temperature >>>> of -20'C to 180'C.
No you haven't. Not using conventional components.
RL
Bugger! I could have sworn it was working at 180'C (along with all the
other parts of the system), but it seems you know better. I must have a
faulty oven.
I'd better warn all the other downhole instrumentation companies too!
But yes, selected conventional components, analog and digital. And yes,
I know that if you extrapolate the graphs, most of the parts de-rate to
negative power dissipation.
We use D2PAK mosfets and lead-free solder, with relow temps around
240c. That works fine.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature
of -20'C to 180'C.
On Wed, 6 Dec 2023 15:26:00 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature
of -20'C to 180'C.
How about reducing the _amplifier_ temperature range with a Peltier
element ?
This will:
- increase component life time
- simplify biasing
However, there are several issues with Peltiers:
- it doesn't tolerate strong vibration
- sufficient extra power must be available to drive the element,
often more than the power that you want to transfer
out from the amplifier
- the hot side can be well above 200 C if the ambient is hot,
remember to dissipate both the amplifier losses and in
addition the power for the Peltier
The Peltier can be used to cool the amplifier in a hot environment.
Reversing the Peltier current and it can be used to warm up the
amplifier in a cold environment. Thus the bias design is simplified.
On 07/12/2023 15:04, legg wrote:
On Wed, 6 Dec 2023 15:26:00 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature >>> of -20'C to 180'C.
No you haven't. Not using conventional components.
RL
Bugger! I could have sworn it was working at 180'C (along with all the
other parts of the system), but it seems you know better. I must have a >faulty oven.
I'd better warn all the other downhole instrumentation companies too!
But yes, selected conventional components, analog and digital. And yes,
I know that if you extrapolate the graphs, most of the parts de-rate to >negative power dissipation.
onsdag den 6. december 2023 kl. 16.27.07 UTC+1 skrev Clive Arthur:
, and if I'd used say a rubber diode to get some quiescent
current, I think it would be very difficult to control Iq well enough
over the temperature range.
tried how bad it is if you bolt bias transistor to TR6 and TR8?
On 08/12/2023 02:45, John Larkin wrote:
On Thu, 7 Dec 2023 15:26:52 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
On 07/12/2023 15:04, legg wrote:
On Wed, 6 Dec 2023 15:26:00 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V >>>>> single supply, and to go up to 100kHz with a working ambient temperature >>>>> of -20'C to 180'C.
No you haven't. Not using conventional components.
RL
Bugger! I could have sworn it was working at 180'C (along with all the
other parts of the system), but it seems you know better. I must have a >>> faulty oven.
I'd better warn all the other downhole instrumentation companies too!
But yes, selected conventional components, analog and digital. And yes, >>> I know that if you extrapolate the graphs, most of the parts de-rate to
negative power dissipation.
We use D2PAK mosfets and lead-free solder, with relow temps around
240c. That works fine.
SiC is good too.
When I first started working in this area, I was very surprised on my
first day to see a colleague doing a crude temperature test using a hot
air gun and a thermocouple, just checking before a long term test in an
oven.
I'm not giving anything away which isn't well known to those in the
business by saying the part in question was an ordinary 8-bit PIC.
Operating at 180'C. An 85'C part IIRC, though I suspect the only
difference between that and a 125'C part is the part number. I've had
PICs running at 200'C, though 180'C is the usual benchmark.
Those involved have lists of components which they've tested. That
takes considerable time and money so they don't readily reveal that >information.
On Thu, 7 Dec 2023 15:26:52 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
On 07/12/2023 15:04, legg wrote:
On Wed, 6 Dec 2023 15:26:00 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V >>>> single supply, and to go up to 100kHz with a working ambient temperature >>>> of -20'C to 180'C.
No you haven't. Not using conventional components.
RL
Bugger! I could have sworn it was working at 180'C (along with all the
other parts of the system), but it seems you know better. I must have a
faulty oven.
I'd better warn all the other downhole instrumentation companies too!
But yes, selected conventional components, analog and digital. And yes,
I know that if you extrapolate the graphs, most of the parts de-rate to
negative power dissipation.
E.M.Cherry's PA circuitry and their clones can form rugged designs,
but they were never intended for data transmission. What's the
format?
Thermal compensation of quiescent biasing was a good trick, then, but
you'd have to reconsider options over an extended range. Claiming to
have done such a design without doing so is silly.
You may have to satisfy yourself with something that doesn't work
very well (or at all) at room temperature, if you stick with the
limitations of Self's variation (intended to address inaudible
distortion).
Self-heating to within the range isn't out of the question, though inconvenient on the bench.
RL
Setting up a temp chamber is a nuisance. I use a cardboard box with
some padding inside, a heat gun, and freeze spray on my bench,
whenever I can.
https://www.dropbox.com/scl/fi/ncxlgwgyvyoxexzmfqyk3/T660_Temp_Chamber.jpg?rlkey=oud1q89ygu5nafd2i5nym6jii&raw=1
On 08/12/2023 15:44, John Larkin wrote:
<snip>
Setting up a temp chamber is a nuisance. I use a cardboard box with
some padding inside, a heat gun, and freeze spray on my bench,
whenever I can.
https://www.dropbox.com/scl/fi/ncxlgwgyvyoxexzmfqyk3/T660_Temp_Chamber.jpg?rlkey=oud1q89ygu5nafd2i5nym6jii&raw=1
Cardboard is good for a short time, I often use similar with a heat gun >blowing in.
A proper lab oven for long-term testing of course. First time I did
this, I thought I'd inspect after 1000 hours. Too enthusiastic - cooled >down, reached in and zap, static, dead board.
I often use a fairly wide-mouthed stainless steel Thermos (Dewar) flask
(my boards are long and thin) with an aquarium pump on the bench pumping
air through an insulated high power resistor (the type with the water
cooling tube down the middle) and put the hot silicone pipe to the
bottom of the flask with wadding in the top.
Cheap Chinese temperature controller and you have something compact and >relatively safe using minimal power.
On 08/12/2023 15:35, legg wrote:
On Thu, 7 Dec 2023 15:26:52 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
On 07/12/2023 15:04, legg wrote:
On Wed, 6 Dec 2023 15:26:00 +0000, Clive Arthur
<clive@nowaytoday.co.uk> wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V >>>>> single supply, and to go up to 100kHz with a working ambient temperature >>>>> of -20'C to 180'C.
No you haven't. Not using conventional components.
RL
Bugger! I could have sworn it was working at 180'C (along with all the
other parts of the system), but it seems you know better. I must have a >>> faulty oven.
I'd better warn all the other downhole instrumentation companies too!
But yes, selected conventional components, analog and digital. And yes, >>> I know that if you extrapolate the graphs, most of the parts de-rate to
negative power dissipation.
E.M.Cherry's PA circuitry and their clones can form rugged designs,
but they were never intended for data transmission. What's the
format?
Thermal compensation of quiescent biasing was a good trick, then, but
you'd have to reconsider options over an extended range. Claiming to
have done such a design without doing so is silly.
You may have to satisfy yourself with something that doesn't work
very well (or at all) at room temperature, if you stick with the
limitations of Self's variation (intended to address inaudible
distortion).
Self-heating to within the range isn't out of the question, though
inconvenient on the bench.
RL
So, first of all you tell me that I haven't done what I have done (and
what many others could have done), then you tell me that it won't work
very well, if at all.
Who's going to break the news to the users? They'll be understandably
upset that their systems work by some magic other than the electronics
they paid for.
It performs well enough to pass the acceptance tests without issue.
What I'm looking towards is the next iteration - can I tweak what I have
or should I start afresh?
The latter is always preferable to the designer, the former to their >paymasters. High temperature work always takes longer and costs more,
mostly because of the testing and the restricted range of components.
On 06/12/2023 15:26, Clive Arthur wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature
of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here...
http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but too
far and you have oscillation. Also, problems occur with slew rate
limiting due to Cdom, the TR5 constant current, and the increase in hFE
of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead. Seems
stable and not slew rate limiting. Took a lot longer to do than that
sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8 probably
get hotter and their Vbe would drop by more than the diodes, so it's
class B. As it runs at a high temperature, I obviously want to reduce
dissipation, and if I'd used say a rubber diode to get some quiescent
current, I think it would be very difficult to control Iq well enough
over the temperature range.
So I have a circuit which works well enough, but could be better with
regard to crossover distortion (though it's lower than I would have
thought). In this application, the better the signal quality, the
higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
Maintaining class AB bias over that temperature range is going to be >difficult. I'd look instead at having no Vbias so the power stage
operates pure class B and then have a fast small class A stage fill in
the cross-over distortion. In other words the Quad feed-forward aka
current dumping idea of the 1970s.
On a sunny day (Wed, 6 Dec 2023 18:54:05 +0000) it happened piglet <erichpwagner@hotmail.com> wrote in <ukqg0e$si84$1@dont-email.me>:
On 06/12/2023 15:26, Clive Arthur wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V
single supply, and to go up to 100kHz with a working ambient temperature >>> of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here...
http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but too
far and you have oscillation. Also, problems occur with slew rate
limiting due to Cdom, the TR5 constant current, and the increase in hFE
of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead. Seems
stable and not slew rate limiting. Took a lot longer to do than that
sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8 probably
get hotter and their Vbe would drop by more than the diodes, so it's
class B. As it runs at a high temperature, I obviously want to reduce
dissipation, and if I'd used say a rubber diode to get some quiescent
current, I think it would be very difficult to control Iq well enough
over the temperature range.
So I have a circuit which works well enough, but could be better with
regard to crossover distortion (though it's lower than I would have
thought). In this application, the better the signal quality, the
higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
Maintaining class AB bias over that temperature range is going to be
difficult. I'd look instead at having no Vbias so the power stage
operates pure class B and then have a fast small class A stage fill in
the cross-over distortion. In other words the Quad feed-forward aka
current dumping idea of the 1970s.
I liked that idea.
On 09/12/2023 6:54 am, Jan Panteltje wrote:
On a sunny day (Wed, 6 Dec 2023 18:54:05 +0000) it happened piglet
<erichpwagner@hotmail.com> wrote in <ukqg0e$si84$1@dont-email.me>:
On 06/12/2023 15:26, Clive Arthur wrote:
I'm not an analog design expert, but needs must.
I recently adapted a Doug Self audio amplifier design for use on a 60V >>>> single supply, and to go up to 100kHz with a working ambient temperature >>>> of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here...
http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but too >>>> far and you have oscillation. Also, problems occur with slew rate
limiting due to Cdom, the TR5 constant current, and the increase in hFE >>>> of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead. Seems >>>> stable and not slew rate limiting. Took a lot longer to do than that >>>> sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8 probably >>>> get hotter and their Vbe would drop by more than the diodes, so it's
class B. As it runs at a high temperature, I obviously want to reduce >>>> dissipation, and if I'd used say a rubber diode to get some quiescent
current, I think it would be very difficult to control Iq well enough
over the temperature range.
So I have a circuit which works well enough, but could be better with
regard to crossover distortion (though it's lower than I would have
thought). In this application, the better the signal quality, the
higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
Maintaining class AB bias over that temperature range is going to be
difficult. I'd look instead at having no Vbias so the power stage
operates pure class B and then have a fast small class A stage fill in
the cross-over distortion. In other words the Quad feed-forward aka
current dumping idea of the 1970s.
I liked that idea.
Yes, the idea appealled to me too. I built the circuit from Quad 405 in:
<https://www.worldradiohistory.com/UK/Wireless-World/70s/Wireless-World-1975-12.pdf>
It was very robust amp worked very well for me.
On Wednesday, December 6, 2023 at 6:22:34 PM UTC-5, Glen Walpert wrote:a4ec-f89b60d40589?type=1&inlineName=True
On Wed, 6 Dec 2023 22:49:40 +0000, Clive Arthur wrote:
On 06/12/2023 21:19, Phil Hobbs wrote:Some years ago Jim Thompson posted an audio amplifier design which used
On 2023-12-06 10:26, Clive Arthur wrote:
I'm not an analog design expert, but needs must.Some local feedback around the output stage would get my vote. The
I recently adapted a Doug Self audio amplifier design for use on a
60V single supply, and to go up to 100kHz with a working ambient
temperature of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here...
http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but
too far and you have oscillation. Also, problems occur with slew
rate limiting due to Cdom, the TR5 constant current, and the
increase in hFE of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead.
Seems stable and not slew rate limiting. Took a lot longer to do
than that sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8
probably get hotter and their Vbe would drop by more than the
diodes, so it's class B. As it runs at a high temperature, I
obviously want to reduce dissipation, and if I'd used say a rubber
diode to get some quiescent current, I think it would be very
difficult to control Iq well enough over the temperature range.
So I have a circuit which works well enough, but could be better
with regard to crossover distortion (though it's lower than I would
have thought). In this application, the better the signal quality,
the higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover
point.
Sziklai pairs have their own local feedback, but that doesn't fix
the crossover problem.
Another approach would be to turn TR4 into a diff pair. TR3's
collector swing is going to waste, and that would let you keep the
open-loop gain the same, while stabilizing the tail current.
Cheers
Phil Hobbs
Thanks, that second one particularly sounds like a good idea. I'll
see what the sim says. Local feedback on the output stage sounds
trickier,
I'll have to think.
One thing I thought of was to use multiple smaller output pairs in
parallel, and have a DC offset for each one. Imagine replacing the
Vbias diode with a string of a few series diodes, and connecting the
bases of one output pair across D1, the next pair across D2 etc.
Would need resistors from each pair of emitters to the output. That
should give lower crossover in more places. Maybe.
current mirrors to provide bias to the output transistors for the
express purpose of keeping crossover distortion low over a large
temperature range. He claimed it was the bees knees, but a quick search
failed to turn it up. Perhaps someone else saved it or remembers the
thread?
This circuit looks like what you're talking about, MC34071 :
https://www.javanelec.com/CustomAjax/GetAppDocument/9feb1d77-b648-447f-
But it's a typical OA input and intermediate gain stage so the gain is
very large. The discrete Self circuit doesn't come close. Without gain
with bw near 10MHz, suppression of large signal output distortion is
going to be kinda weak.
That may use the same bias method, but I was thinking of a discrete transistor design, posted on his web site and possibly still available somewhere:
From: Jim Thompson <To-Email-Use-The-Envelope-Icon@On-My-Web-Site.com> Newsgroups: sci.electronics.design
Subject: Unusual Bias Method
Date: Sun, 24 Nov 2013 08:53:30 -0700
Message-ID: <sl749997np4gq2giqk9d8k77orheh6qt3d@4ax
Here's half of the full H-bridge amplifiers that I built for my 1977
280Z... Image scanned in quarters and pieced together for easier understanding...
<http://www.analog-innovations.com/SED/Half_Bridge_for_77_280Z.pdf>
No one has commented on the unusual bias scheme in this amplifier since I originally posted it.
No actual circuit designers in our midst ?:-}
...Jim Thompson
Glen Walpert <nospam@null.void> wrote:
That may use the same bias method, but I was thinking of a discrete
transistor design, posted on his web site and possibly still available
somewhere:
From: Jim Thompson <To-Email-Use-The-Envelope-Icon@On-My-Web-Site.com>
Newsgroups: sci.electronics.design
Subject: Unusual Bias Method
Date: Sun, 24 Nov 2013 08:53:30 -0700
Message-ID: <sl749997np4gq2giqk9d8k77orheh6qt3d@4ax
Here's half of the full H-bridge amplifiers that I built for my 1977
280Z... Image scanned in quarters and pieced together for easier
understanding...
<http://www.analog-innovations.com/SED/Half_Bridge_for_77_280Z.pdf>
No one has commented on the unusual bias scheme in this amplifier since
I originally posted it.
No actual circuit designers in our midst ?:-}
...Jim Thompson
Link times out. Doesn't exist. I seem to recall Phil Hobbs copied Jim's
web site before it was taken down.
On Wed, 6 Dec 2023 22:49:40 +0000, Clive Arthur wrote:
On 06/12/2023 21:19, Phil Hobbs wrote:
On 2023-12-06 10:26, Clive Arthur wrote:
I'm not an analog design expert, but needs must.Some local feedback around the output stage would get my vote. The
I recently adapted a Doug Self audio amplifier design for use on a 60V >>>> single supply, and to go up to 100kHz with a working ambient
temperature of -20'C to 180'C.
I can't show the circuit, but it was based on Fig 1a here...
http://www.douglas-self.com/ampins/dipa/dipa.htm
Cdom needs to come down for this higher frequency application, but too >>>> far and you have oscillation. Also, problems occur with slew rate
limiting due to Cdom, the TR5 constant current, and the increase in
hFE of TR4 with temperature.
So I added an emitter degeneration resistor to TR4 to tame the hFE
variation, removed Cdom and put a smaller C across Rf1 instead. Seems >>>> stable and not slew rate limiting. Took a lot longer to do than that
sentence might imply.
However.
The Vbias is a single diode, can't risk 2 diodes as TR6 & TR8 probably >>>> get hotter and their Vbe would drop by more than the diodes, so it's
class B. As it runs at a high temperature, I obviously want to reduce >>>> dissipation, and if I'd used say a rubber diode to get some quiescent
current, I think it would be very difficult to control Iq well enough
over the temperature range.
So I have a circuit which works well enough, but could be better with
regard to crossover distortion (though it's lower than I would have
thought). In this application, the better the signal quality, the
higher the data rate.
Any ideas for improving crossover distortion, bearing in mind the
temperature range? The signal is OFDM, so pretty much a load of
'random' steps, some of which may be small and at a crossover point.
Sziklai pairs have their own local feedback, but that doesn't fix the
crossover problem.
Another approach would be to turn TR4 into a diff pair. TR3's
collector swing is going to waste, and that would let you keep the
open-loop gain the same, while stabilizing the tail current.
Cheers
Phil Hobbs
Thanks, that second one particularly sounds like a good idea. I'll see
what the sim says. Local feedback on the output stage sounds trickier,
I'll have to think.
One thing I thought of was to use multiple smaller output pairs in
parallel, and have a DC offset for each one. Imagine replacing the
Vbias diode with a string of a few series diodes, and connecting the
bases of one output pair across D1, the next pair across D2 etc. Would
need resistors from each pair of emitters to the output. That should
give lower crossover in more places. Maybe.
Some years ago Jim Thompson posted an audio amplifier design which used >current mirrors to provide bias to the output transistors for the express >purpose of keeping crossover distortion low over a large temperature
range. He claimed it was the bees knees, but a quick search failed to
turn it up. Perhaps someone else saved it or remembers the thread?
Glen
On Wed, 06 Dec 2023 23:22:26 GMT, Glen Walpert <nospam@null.void>
wrote:
Some years ago Jim Thompson posted an audio amplifier design which used
current mirrors to provide bias to the output transistors for the express
purpose of keeping crossover distortion low over a large temperature
range. He claimed it was the bees knees, but a quick search failed to
turn it up. Perhaps someone else saved it or remembers the thread?
Glen
From old LTSpice trash here;
http://ve3ute.ca/query/Half_Bridge_for_77_280Z_thompson.pdf
RL
On 11/12/2023 14:05, legg wrote:
On Wed, 06 Dec 2023 23:22:26 GMT, Glen Walpert <nospam@null.void>
wrote:
<snip>
Some years ago Jim Thompson posted an audio amplifier design which used
current mirrors to provide bias to the output transistors for the express >>> purpose of keeping crossover distortion low over a large temperature
range. He claimed it was the bees knees, but a quick search failed to
turn it up. Perhaps someone else saved it or remembers the thread?
Glen
From old LTSpice trash here;
http://ve3ute.ca/query/Half_Bridge_for_77_280Z_thompson.pdf
RL
Thanks!
Looks like the top output Darlington is AC coupled and when the
comparator detects a quiescent current through the output resistors >transitioning to less than some value, it pumps the upper Darlington
base voltage up a bit, otherwise, the upper Darlington base voltage
drifts down.
Is that about right?
Not sure it would work in my application as my signal isn't continuous -
it spends some proportion of the time idling at half supply. Still, I
could probably arrange a clock to force a comparator sample somehow.
Or maybe make the adjustment non-volatile (digipot?) and clock it both
up and down. The signal comes from a DAC, so I do have access to timing >signals.
On 11/12/2023 14:05, legg wrote:
On Wed, 06 Dec 2023 23:22:26 GMT, Glen Walpert <nospam@null.void>
wrote:
<snip>
Some years ago Jim Thompson posted an audio amplifier design which
used current mirrors to provide bias to the output transistors for the
express purpose of keeping crossover distortion low over a large
temperature range. He claimed it was the bees knees, but a quick
search failed to turn it up. Perhaps someone else saved it or
remembers the thread?
Glen
From old LTSpice trash here;
http://ve3ute.ca/query/Half_Bridge_for_77_280Z_thompson.pdf
RL
Thanks!
Looks like the top output Darlington is AC coupled and when the
comparator detects a quiescent current through the output resistors transitioning to less than some value, it pumps the upper Darlington
base voltage up a bit, otherwise, the upper Darlington base voltage
drifts down.
Is that about right?
If both output transistors are briefly off or very nearly off while the >output is increasing through crossover (zero or near zero current through >both emitter resistors), then the LM311 goes high due to the lag through
the RC on the negative input, delivering additional current to the
current mirror with illegible designations through D1, pulling current
from the 20uF 10V capacitor, increasing it's voltage thus increasing the
bias offset provided by Q5 and Q6 until there is enough bias voltage >difference to insure some small overlap in the on time of the output >transistors. Q1 and Q2 appear to keep the bias voltages centered between
the rails, and possibly Q8 pulls the negative input of the comparator
down enough to prevent noise from turning it on with no input?. (Not at
all sure about Q8, it might do more than that).
Am I close? Hints on Q8?
Regards,
Glen
What's the SPICE quiescent bias? Back of the envelope, I get 75 or 80mA.
How do you pick R15/R16/C4? Looks like it's to bootstrap the bias above
the 13.3V rail with a time constant longer than the roll-off of the >amplifier.
Best regards,
Spehro Pefhany
Latest version....Bias_Amplifier.zip>
<http://www.analog-innovations.com/SED/JimThompsons_A-B-
On Monday, December 11, 2023 at 10:50:04 AM UTC-5, legg wrote:
On Mon, 11 Dec 2023 15:01:25 +0000, Clive Arthur
<cl...@nowaytoday.co.uk> wrote:
On 11/12/2023 14:05, legg wrote:A lot of the bumph is dedicated only to biasing and it would take
On Wed, 06 Dec 2023 23:22:26 GMT, Glen Walpert
<nos...@null.void> wrote:
<snip>
Some years ago Jim Thompson posted an audio amplifier design
which used current mirrors to provide bias to the output
transistors for the express purpose of keeping crossover
distortion low over a large temperature range. He claimed it
was the bees knees, but a quick search failed to turn it up.
Perhaps someone else saved it or remembers the thread?
Glen
From old LTSpice trash here;
http://ve3ute.ca/query/Half_Bridge_for_77_280Z_thompson.pdf
RL
Thanks!
Looks like the top output Darlington is AC coupled and when the
comparator detects a quiescent current through the output
resistors transitioning to less than some value, it pumps the
upper Darlington base voltage up a bit, otherwise, the upper
Darlington base voltage drifts down.
Is that about right?
Not sure it would work in my application as my signal isn't
continuous - it spends some proportion of the time idling at half
supply. Still, I could probably arrange a clock to force a
comparator sample somehow.
Or maybe make the adjustment non-volatile (digipot?) and clock it
both up and down. The signal comes from a DAC, so I do have
access to timing signals.
some doing to get it to work over temperature given those
polarized cap sizes. Integrated darlingtons are also best avoided.
By 'wide range', the author was talking standard industrial
temperatures.
You'd also have to do some thin'in around the gain-setting regime.
Doubt this was a consideration in this drawing ( . . . 'or' . .
.), nor was 100KHz ( hence zobel network ).
I don't see quiescent conditions being an issue, but start-up and
shutdown could be surprising. Not sure that was Thompson's strong
point.
RL
That 100u base-to-base bias cap is probably needed to dominant poll
stabilize the LM311 amp more than anything else.
On 2023-12-11 12:09, Fred Bloggs wrote:
On Monday, December 11, 2023 at 10:50:04?AM UTC-5, legg wrote:
On Mon, 11 Dec 2023 15:01:25 +0000, Clive Arthur
<cl...@nowaytoday.co.uk> wrote:
On 11/12/2023 14:05, legg wrote:A lot of the bumph is dedicated only to biasing and it would take
On Wed, 06 Dec 2023 23:22:26 GMT, Glen Walpert
<nos...@null.void> wrote:
<snip>
Some years ago Jim Thompson posted an audio amplifier design
which used current mirrors to provide bias to the output
transistors for the express purpose of keeping crossover
distortion low over a large temperature range. He claimed it
was the bees knees, but a quick search failed to turn it up.
Perhaps someone else saved it or remembers the thread?
Glen
From old LTSpice trash here;
http://ve3ute.ca/query/Half_Bridge_for_77_280Z_thompson.pdf
RL
Thanks!
Looks like the top output Darlington is AC coupled and when the
comparator detects a quiescent current through the output
resistors transitioning to less than some value, it pumps the
upper Darlington base voltage up a bit, otherwise, the upper
Darlington base voltage drifts down.
Is that about right?
Not sure it would work in my application as my signal isn't
continuous - it spends some proportion of the time idling at half
supply. Still, I could probably arrange a clock to force a
comparator sample somehow.
Or maybe make the adjustment non-volatile (digipot?) and clock it
both up and down. The signal comes from a DAC, so I do have
access to timing signals.
some doing to get it to work over temperature given those
polarized cap sizes. Integrated darlingtons are also best avoided.
By 'wide range', the author was talking standard industrial
temperatures.
You'd also have to do some thin'in around the gain-setting regime.
Doubt this was a consideration in this drawing ( . . . 'or' . .
.), nor was 100KHz ( hence zobel network ).
I don't see quiescent conditions being an issue, but start-up and
shutdown could be surprising. Not sure that was Thompson's strong
point.
RL
That 100u base-to-base bias cap is probably needed to dominant poll
stabilize the LM311 amp more than anything else.
I'm not sure that we're looking at the same schematic. In the one I
have, "Half_Bridge_for_77_280Z_thompson.pdf", the LM311 is a weird sort
of switching bias supervisor, not a linear amp. It's running as a
normal open-collector comparator, with its output wire-ORed with the
shutdown transistor Q7.
When it fires, or the shutdown line is high, it steals Q3's base bias.
That causes Darlington Q5/Q6 to turn on more, with a TC of about 2
seconds. That reduces the quiescent bias.
In small-signal conditions, that'll just oscillate irregularly and keep
the class-A bias current of very roughly 60 mA. (*) In large-signal >conditions, the comparator will be pulling low most of the time, which >reduces the quiescent bias progressively. (If the gain of the bias loop
is high enough, it may not drift that far, but I'd probably need to use
SPICE to find that out.)
When the shutdown pin is active, the class-A bias will gradually go to
0, turning the output totem pole into a really bad class B. (One
gathers that the shutdown turns off the audio input as well.)
One aspect that I don't understand well is Q3. With a 3k/100 ohm
voltage divider in the Q4 leg, ISTM that the base voltage of Q3 will be >nearly the same as that of Q2, which is driven from a 620 ohm / 22 ohm >divider (plus various V_BEs). That makes the current through the 220
ohm hard to estimate by eyeball. (The average current is obviously
going to be small, on account of that 100k resistor.)
(I sort of gather that it's pretty tweaky, due to that scribbled-in 10k
to ground from the bases of Q3 and Q4.)
Cheers
Phil Hobbs
(*) That 60 mA number is based on the ~20 mA emitter current of Q1.
That'll drop about 450 mV across the 22 ohms, which translates to about
1.5 mA collector current in Q8. That puts roughly 40 mV across the 27
ohms, which divided by 0.66 ohms gets you roughly 60 mA of Class-A bias >current.
On Mon, 11 Dec 2023 18:05:46 -0500, Phil Hobbs <pcdhSpamMeSenseless@electrooptical.net> wrote:
On 2023-12-11 12:09, Fred Bloggs wrote:
On Monday, December 11, 2023 at 10:50:04?AM UTC-5, legg wrote:
On Mon, 11 Dec 2023 15:01:25 +0000, Clive Arthur
<cl...@nowaytoday.co.uk> wrote:
On 11/12/2023 14:05, legg wrote:A lot of the bumph is dedicated only to biasing and it would take
On Wed, 06 Dec 2023 23:22:26 GMT, Glen Walpert
<nos...@null.void> wrote:
<snip>
Some years ago Jim Thompson posted an audio amplifier design
which used current mirrors to provide bias to the output
transistors for the express purpose of keeping crossover
distortion low over a large temperature range. He claimed it
was the bees knees, but a quick search failed to turn it up.
Perhaps someone else saved it or remembers the thread?
Glen
From old LTSpice trash here;
http://ve3ute.ca/query/Half_Bridge_for_77_280Z_thompson.pdf
RL
Thanks!
Looks like the top output Darlington is AC coupled and when the
comparator detects a quiescent current through the output
resistors transitioning to less than some value, it pumps the
upper Darlington base voltage up a bit, otherwise, the upper
Darlington base voltage drifts down.
Is that about right?
Not sure it would work in my application as my signal isn't
continuous - it spends some proportion of the time idling at half
supply. Still, I could probably arrange a clock to force a
comparator sample somehow.
Or maybe make the adjustment non-volatile (digipot?) and clock it
both up and down. The signal comes from a DAC, so I do have
access to timing signals.
some doing to get it to work over temperature given those
polarized cap sizes. Integrated darlingtons are also best avoided.
By 'wide range', the author was talking standard industrial
temperatures.
You'd also have to do some thin'in around the gain-setting regime.
Doubt this was a consideration in this drawing ( . . . 'or' . .
.), nor was 100KHz ( hence zobel network ).
I don't see quiescent conditions being an issue, but start-up and
shutdown could be surprising. Not sure that was Thompson's strong
point.
RL
That 100u base-to-base bias cap is probably needed to dominant poll
stabilize the LM311 amp more than anything else.
I'm not sure that we're looking at the same schematic. In the one I
have, "Half_Bridge_for_77_280Z_thompson.pdf", the LM311 is a weird sort
of switching bias supervisor, not a linear amp. It's running as a
normal open-collector comparator, with its output wire-ORed with the
shutdown transistor Q7.
When it fires, or the shutdown line is high, it steals Q3's base bias.
That causes Darlington Q5/Q6 to turn on more, with a TC of about 2
seconds. That reduces the quiescent bias.
In small-signal conditions, that'll just oscillate irregularly and keep
the class-A bias current of very roughly 60 mA. (*) In large-signal
conditions, the comparator will be pulling low most of the time, which
reduces the quiescent bias progressively. (If the gain of the bias loop
is high enough, it may not drift that far, but I'd probably need to use
SPICE to find that out.)
When the shutdown pin is active, the class-A bias will gradually go to
0, turning the output totem pole into a really bad class B. (One
gathers that the shutdown turns off the audio input as well.)
One aspect that I don't understand well is Q3. With a 3k/100 ohm
voltage divider in the Q4 leg, ISTM that the base voltage of Q3 will be
nearly the same as that of Q2, which is driven from a 620 ohm / 22 ohm
divider (plus various V_BEs). That makes the current through the 220
ohm hard to estimate by eyeball. (The average current is obviously
going to be small, on account of that 100k resistor.)
(I sort of gather that it's pretty tweaky, due to that scribbled-in 10k
to ground from the bases of Q3 and Q4.)
Cheers
Phil Hobbs
(*) That 60 mA number is based on the ~20 mA emitter current of Q1.
That'll drop about 450 mV across the 22 ohms, which translates to about
1.5 mA collector current in Q8. That puts roughly 40 mV across the 27
ohms, which divided by 0.66 ohms gets you roughly 60 mA of Class-A bias
current.
Phil,
The 'bumph' controlling the biasing is a sub-audible current switch
into fairly large capacitors, so simulation would have to take this
into account.
In my internet/bugs/ampjt folder there are a number of simulations
posted by Jim around 2013. They used the .op spice directive to
establish DC operating point values only.
I couldn't get it to operate as an amplifier (starts with static
latched-off biasing) using his TL081/071 subcircuit. Using the
basic LTSpice single or doublepole OA would allow it to demonstrate
a signal path in a .tran simulation.
His sims left out the electrolytic cap on the base of Q2 and the
gain was set to simple unity (inverting) using 10K resistors
and a cap-coupled source.
Crossover distortion was easily visible. As I couldn't see any
slow-moving voltages or currents in the biasing section to
correct this, while a signal was being processed, or any reason
why they should change (with the LM311 inputs overloaded by
normal operating current), I set the thing aside.
RL
In 1977 it wasn't that easy to do a fully-differential measurement of a
40-mV signal sitting on 12 Vpp of audio.
Re-framing the problem as preventing the measured voltage from falling
much below 40 mV, and letting it gradually decrease otherwise, is an interesting approach.
On 2023-12-12 09:17, legg wrote:<snip>
On Mon, 11 Dec 2023 18:05:46 -0500, Phil Hobbs
<pcdhSpamMeSenseless@electrooptical.net> wrote:
On 2023-12-11 12:09, Fred Bloggs wrote:
One aspect that I don't understand well is Q3. With a 3k/100 ohm
voltage divider in the Q4 leg, ISTM that the base voltage of Q3 will be
nearly the same as that of Q2, which is driven from a 620 ohm / 22 ohm
divider (plus various V_BEs). That makes the current through the 220
ohm hard to estimate by eyeball. (The average current is obviously
going to be small, on account of that 100k resistor.)
(I sort of gather that it's pretty tweaky, due to that scribbled-in 10k
to ground from the bases of Q3 and Q4.)
Cheers
Phil Hobbs
(*) That 60 mA number is based on the ~20 mA emitter current of Q1.
That'll drop about 450 mV across the 22 ohms, which translates to about
1.5 mA collector current in Q8. That puts roughly 40 mV across the 27
ohms, which divided by 0.66 ohms gets you roughly 60 mA of Class-A bias
current.
Phil,
The 'bumph' controlling the biasing is a sub-audible current switch
into fairly large capacitors, so simulation would have to take this
into account.
In my internet/bugs/ampjt folder there are a number of simulations
posted by Jim around 2013. They used the .op spice directive to
establish DC operating point values only.
I couldn't get it to operate as an amplifier (starts with static
latched-off biasing) using his TL081/071 subcircuit. Using the
basic LTSpice single or doublepole OA would allow it to demonstrate
a signal path in a .tran simulation.
His sims left out the electrolytic cap on the base of Q2 and the
gain was set to simple unity (inverting) using 10K resistors
and a cap-coupled source.
Crossover distortion was easily visible. As I couldn't see any
slow-moving voltages or currents in the biasing section to
correct this, while a signal was being processed, or any reason
why they should change (with the LM311 inputs overloaded by
normal operating current), I set the thing aside.
RL
In 1977 it wasn't that easy to do a fully-differential measurement of a
40-mV signal sitting on 12 Vpp of audio.
Re-framing the problem as preventing the measured voltage from falling
much below 40 mV, and letting it gradually decrease otherwise, is an >interesting approach.
On 12/12/2023 23:43, Phil Hobbs wrote:
<snip>
In 1977 it wasn't that easy to do a fully-differential measurement of a
40-mV signal sitting on 12 Vpp of audio.
Re-framing the problem as preventing the measured voltage from falling
much below 40 mV, and letting it gradually decrease otherwise, is an
interesting approach.
Yes, it's a good idea. I'd go for a current monitor - eg INA169 - to
get things down to the 5V domain (I have a 60V supply) and take it from >there.
On 2023-12-12 09:17, legg wrote:<snip>
On Mon, 11 Dec 2023 18:05:46 -0500, Phil Hobbs
<pcdhSpamMeSenseless@electrooptical.net> wrote:
On 2023-12-11 12:09, Fred Bloggs wrote:
I'm not sure that we're looking at the same schematic. In the one I
have, "Half_Bridge_for_77_280Z_thompson.pdf", the LM311 is a weird sort
of switching bias supervisor, not a linear amp. It's running as a
normal open-collector comparator, with its output wire-ORed with the
shutdown transistor Q7.
When it fires, or the shutdown line is high, it steals Q3's base bias.
That causes Darlington Q5/Q6 to turn on more, with a TC of about 2
seconds. That reduces the quiescent bias.
In small-signal conditions, that'll just oscillate irregularly and keep
the class-A bias current of very roughly 60 mA. (*) In large-signal
conditions, the comparator will be pulling low most of the time, which
reduces the quiescent bias progressively. (If the gain of the bias loop >>> is high enough, it may not drift that far, but I'd probably need to use
SPICE to find that out.)
When the shutdown pin is active, the class-A bias will gradually go to
0, turning the output totem pole into a really bad class B. (One
gathers that the shutdown turns off the audio input as well.)
One aspect that I don't understand well is Q3. With a 3k/100 ohm
voltage divider in the Q4 leg, ISTM that the base voltage of Q3 will be
nearly the same as that of Q2, which is driven from a 620 ohm / 22 ohm
divider (plus various V_BEs). That makes the current through the 220
ohm hard to estimate by eyeball. (The average current is obviously
going to be small, on account of that 100k resistor.)
(I sort of gather that it's pretty tweaky, due to that scribbled-in 10k
to ground from the bases of Q3 and Q4.)
Cheers
Phil Hobbs
(*) That 60 mA number is based on the ~20 mA emitter current of Q1.
That'll drop about 450 mV across the 22 ohms, which translates to about
1.5 mA collector current in Q8. That puts roughly 40 mV across the 27
ohms, which divided by 0.66 ohms gets you roughly 60 mA of Class-A bias
current.
Phil,
The 'bumph' controlling the biasing is a sub-audible current switch
into fairly large capacitors, so simulation would have to take this
into account.
In my internet/bugs/ampjt folder there are a number of simulations
posted by Jim around 2013. They used the .op spice directive to
establish DC operating point values only.
I couldn't get it to operate as an amplifier (starts with static
latched-off biasing) using his TL081/071 subcircuit. Using the
basic LTSpice single or doublepole OA would allow it to demonstrate
a signal path in a .tran simulation.
His sims left out the electrolytic cap on the base of Q2 and the
gain was set to simple unity (inverting) using 10K resistors
and a cap-coupled source.
Crossover distortion was easily visible. As I couldn't see any
slow-moving voltages or currents in the biasing section to
correct this, while a signal was being processed, or any reason
why they should change (with the LM311 inputs overloaded by
normal operating current), I set the thing aside.
RL
In 1977 it wasn't that easy to do a fully-differential measurement of a
40-mV signal sitting on 12 Vpp of audio.
Re-framing the problem as preventing the measured voltage from falling
much below 40 mV, and letting it gradually decrease otherwise, is an >interesting approach.
JT was a smart guy, even if he was a tiny bit too aware of that. ;)
May God hold him in memory eternal.
Cheers
Phil Hobbs
On Tue, 12 Dec 2023 18:43:44 -0500, Phil Hobbs <pcdhSpamMeSenseless@electrooptical.net> wrote:
On 2023-12-12 09:17, legg wrote:<snip>
On Mon, 11 Dec 2023 18:05:46 -0500, Phil Hobbs
<pcdhSpamMeSenseless@electrooptical.net> wrote:
On 2023-12-11 12:09, Fred Bloggs wrote:
I'm not sure that we're looking at the same schematic. In the one I
have, "Half_Bridge_for_77_280Z_thompson.pdf", the LM311 is a weird sort >>>> of switching bias supervisor, not a linear amp. It's running as a
normal open-collector comparator, with its output wire-ORed with the
shutdown transistor Q7.
When it fires, or the shutdown line is high, it steals Q3's base bias. >>>> That causes Darlington Q5/Q6 to turn on more, with a TC of about 2
seconds. That reduces the quiescent bias.
In small-signal conditions, that'll just oscillate irregularly and keep >>>> the class-A bias current of very roughly 60 mA. (*) In large-signal
conditions, the comparator will be pulling low most of the time, which >>>> reduces the quiescent bias progressively. (If the gain of the bias loop >>>> is high enough, it may not drift that far, but I'd probably need to use >>>> SPICE to find that out.)
When the shutdown pin is active, the class-A bias will gradually go to >>>> 0, turning the output totem pole into a really bad class B. (One
gathers that the shutdown turns off the audio input as well.)
One aspect that I don't understand well is Q3. With a 3k/100 ohm
voltage divider in the Q4 leg, ISTM that the base voltage of Q3 will be >>>> nearly the same as that of Q2, which is driven from a 620 ohm / 22 ohm >>>> divider (plus various V_BEs). That makes the current through the 220
ohm hard to estimate by eyeball. (The average current is obviously
going to be small, on account of that 100k resistor.)
(I sort of gather that it's pretty tweaky, due to that scribbled-in 10k >>>> to ground from the bases of Q3 and Q4.)
Cheers
Phil Hobbs
(*) That 60 mA number is based on the ~20 mA emitter current of Q1.
That'll drop about 450 mV across the 22 ohms, which translates to about >>>> 1.5 mA collector current in Q8. That puts roughly 40 mV across the 27 >>>> ohms, which divided by 0.66 ohms gets you roughly 60 mA of Class-A bias >>>> current.
Phil,
The 'bumph' controlling the biasing is a sub-audible current switch
into fairly large capacitors, so simulation would have to take this
into account.
In my internet/bugs/ampjt folder there are a number of simulations
posted by Jim around 2013. They used the .op spice directive to
establish DC operating point values only.
I couldn't get it to operate as an amplifier (starts with static
latched-off biasing) using his TL081/071 subcircuit. Using the
basic LTSpice single or doublepole OA would allow it to demonstrate
a signal path in a .tran simulation.
His sims left out the electrolytic cap on the base of Q2 and the
gain was set to simple unity (inverting) using 10K resistors
and a cap-coupled source.
Crossover distortion was easily visible. As I couldn't see any
slow-moving voltages or currents in the biasing section to
correct this, while a signal was being processed, or any reason
why they should change (with the LM311 inputs overloaded by
normal operating current), I set the thing aside.
RL
In 1977 it wasn't that easy to do a fully-differential measurement of a
40-mV signal sitting on 12 Vpp of audio.
Re-framing the problem as preventing the measured voltage from falling
much below 40 mV, and letting it gradually decrease otherwise, is an
interesting approach.
JT was a smart guy, even if he was a tiny bit too aware of that. ;)
May God hold him in memory eternal.
Cheers
Phil Hobbs
Instead of just bitching about the sims, I replaced both the TL071
subckt and the LM311 subckt with a default OA and an LT1011 from
the standard LTspice library so that the thing runs.
( Be sure to remove or terminate the old subcircuits to ensure
easy sim startup.)
This ran with low crossover distortion at 100KHz and about 50mA
bias current at xover. The bias current could be raised or lowered proportionally through comparator input adjustment.
No big OA slewing requirements.
So the concept could be demonstrated ~accurately in LTSpice.
RL
On Tue, 12 Dec 2023 18:43:44 -0500, Phil Hobbs <pcdhSpamMeSenseless@electrooptical.net> wrote:
On 2023-12-12 09:17, legg wrote:<snip>
On Mon, 11 Dec 2023 18:05:46 -0500, Phil Hobbs
<pcdhSpamMeSenseless@electrooptical.net> wrote:
On 2023-12-11 12:09, Fred Bloggs wrote:
I'm not sure that we're looking at the same schematic. In the one I
have, "Half_Bridge_for_77_280Z_thompson.pdf", the LM311 is a weird sort >>>> of switching bias supervisor, not a linear amp. It's running as a
normal open-collector comparator, with its output wire-ORed with the
shutdown transistor Q7.
When it fires, or the shutdown line is high, it steals Q3's base bias. >>>> That causes Darlington Q5/Q6 to turn on more, with a TC of about 2
seconds. That reduces the quiescent bias.
In small-signal conditions, that'll just oscillate irregularly and keep >>>> the class-A bias current of very roughly 60 mA. (*) In large-signal
conditions, the comparator will be pulling low most of the time, which >>>> reduces the quiescent bias progressively. (If the gain of the bias loop >>>> is high enough, it may not drift that far, but I'd probably need to use >>>> SPICE to find that out.)
When the shutdown pin is active, the class-A bias will gradually go to >>>> 0, turning the output totem pole into a really bad class B. (One
gathers that the shutdown turns off the audio input as well.)
One aspect that I don't understand well is Q3. With a 3k/100 ohm
voltage divider in the Q4 leg, ISTM that the base voltage of Q3 will be >>>> nearly the same as that of Q2, which is driven from a 620 ohm / 22 ohm >>>> divider (plus various V_BEs). That makes the current through the 220
ohm hard to estimate by eyeball. (The average current is obviously
going to be small, on account of that 100k resistor.)
(I sort of gather that it's pretty tweaky, due to that scribbled-in 10k >>>> to ground from the bases of Q3 and Q4.)
Cheers
Phil Hobbs
(*) That 60 mA number is based on the ~20 mA emitter current of Q1.
That'll drop about 450 mV across the 22 ohms, which translates to about >>>> 1.5 mA collector current in Q8. That puts roughly 40 mV across the 27 >>>> ohms, which divided by 0.66 ohms gets you roughly 60 mA of Class-A bias >>>> current.
Phil,
The 'bumph' controlling the biasing is a sub-audible current switch
into fairly large capacitors, so simulation would have to take this
into account.
In my internet/bugs/ampjt folder there are a number of simulations
posted by Jim around 2013. They used the .op spice directive to
establish DC operating point values only.
I couldn't get it to operate as an amplifier (starts with static
latched-off biasing) using his TL081/071 subcircuit. Using the
basic LTSpice single or doublepole OA would allow it to demonstrate
a signal path in a .tran simulation.
His sims left out the electrolytic cap on the base of Q2 and the
gain was set to simple unity (inverting) using 10K resistors
and a cap-coupled source.
Crossover distortion was easily visible. As I couldn't see any
slow-moving voltages or currents in the biasing section to
correct this, while a signal was being processed, or any reason
why they should change (with the LM311 inputs overloaded by
normal operating current), I set the thing aside.
RL
In 1977 it wasn't that easy to do a fully-differential measurement of a
40-mV signal sitting on 12 Vpp of audio.
Re-framing the problem as preventing the measured voltage from falling
much below 40 mV, and letting it gradually decrease otherwise, is an
interesting approach.
JT was a smart guy, even if he was a tiny bit too aware of that. ;)
May God hold him in memory eternal.
Cheers
Phil Hobbs
Instead of just bitching about the sims, I replaced both the TL071
subckt and the LM311 subckt with a default OA and an LT1011 from
the standard LTspice library so that the thing runs.
( Be sure to remove or terminate the old subcircuits to ensure
easy sim startup.)
This ran with low crossover distortion at 100KHz and about 50mA
bias current at xover. The bias current could be raised or lowered proportionally through comparator input adjustment.
No big OA slewing requirements.
So the concept could be demonstrated ~accurately in LTSpice.
RL
On 2023-12-14 14:32, legg wrote:<snip>
On Tue, 12 Dec 2023 18:43:44 -0500, Phil Hobbs
<pcdhSpamMeSenseless@electrooptical.net> wrote:
Instead of just bitching about the sims, I replaced both the TL071
subckt and the LM311 subckt with a default OA and an LT1011 from
the standard LTspice library so that the thing runs.
( Be sure to remove or terminate the old subcircuits to ensure
easy sim startup.)
This ran with low crossover distortion at 100KHz and about 50mA
bias current at xover. The bias current could be raised or lowered
proportionally through comparator input adjustment.
No big OA slewing requirements.
So the concept could be demonstrated ~accurately in LTSpice.
RL
Yeah, I did it too, using 2N3904s and one 2N3906 for the small signal
stuff, D44H11/D45H11 for the output stages of the Darlingtons, a >UniversalOpAmp2 configured to look like an LF356 (4 MHz, 12 V/us), an
RH111 (rad hard comparator from the LTspice 17.1 library) and a 4-ohm
load.
Looks pretty good, for a car amp of that vintage. The bias circuit
works well, anyway.
Cheers
Phil Hobbs
On Monday, December 11, 2023 at 6:06:08 PM UTC-5, Phil Hobbs wrote:RL
On 2023-12-11 12:09, Fred Bloggs wrote:
On Monday, December 11, 2023 at 10:50:04 AM UTC-5, legg wrote:
On Mon, 11 Dec 2023 15:01:25 +0000, Clive Arthur
<cl...@nowaytoday.co.uk> wrote:
On 11/12/2023 14:05, legg wrote:
On Wed, 06 Dec 2023 23:22:26 GMT, Glen Walpert
<nos...@null.void> wrote:
<snip>
Some years ago Jim Thompson posted an audio amplifier
design which used current mirrors to provide bias to the
output transistors for the express purpose of keeping
crossover distortion low over a large temperature range.
He claimed it was the bees knees, but a quick search
failed to turn it up. Perhaps someone else saved it or
remembers the thread?
Glen
From old LTSpice trash here;
http://ve3ute.ca/query/Half_Bridge_for_77_280Z_thompson.pdf
I'm not sure that we're looking at the same schematic. In the oneA lot of the bumph is dedicated only to biasing and it would
Thanks!
Looks like the top output Darlington is AC coupled and when
the comparator detects a quiescent current through the
output resistors transitioning to less than some value, it
pumps the upper Darlington base voltage up a bit, otherwise,
the upper Darlington base voltage drifts down.
Is that about right?
Not sure it would work in my application as my signal isn't
continuous - it spends some proportion of the time idling at
half supply. Still, I could probably arrange a clock to force
a comparator sample somehow.
Or maybe make the adjustment non-volatile (digipot?) and
clock it both up and down. The signal comes from a DAC, so I
do have access to timing signals.
take some doing to get it to work over temperature given those
polarized cap sizes. Integrated darlingtons are also best
avoided. By 'wide range', the author was talking standard
industrial temperatures.
You'd also have to do some thin'in around the gain-setting
regime. Doubt this was a consideration in this drawing ( . . .
'or' . . .), nor was 100KHz ( hence zobel network ).
I don't see quiescent conditions being an issue, but start-up
and shutdown could be surprising. Not sure that was Thompson's
strong point.
RL
That 100u base-to-base bias cap is probably needed to dominant
poll stabilize the LM311 amp more than anything else.
I have, "Half_Bridge_for_77_280Z_thompson.pdf", the LM311 is a
weird sort of switching bias supervisor, not a linear amp. It's
running as a normal open-collector comparator, with its output
wire-ORed with the shutdown transistor Q7.
That's crazy. For one thing, audio speakers do not respond well to discontinuities in the drive voltage. They do things like snap,
crackle and pop.
When it fires, or the shutdown line is high, it steals Q3's base
bias. That causes Darlington Q5/Q6 to turn on more, with a TC of
about 2 seconds. That reduces the quiescent bias.
In small-signal conditions, that'll just oscillate irregularly and
keep the class-A bias current of very roughly 60 mA. (*) In
large-signal conditions, the comparator will be pulling low most of
the time, which reduces the quiescent bias progressively. (If the
gain of the bias loop is high enough, it may not drift that far,
but I'd probably need to use SPICE to find that out.)
When the shutdown pin is active, the class-A bias will gradually go
to 0, turning the output totem pole into a really bad class B.
(One gathers that the shutdown turns off the audio input as well.)
One aspect that I don't understand well is Q3. With a 3k/100 ohm
voltage divider in the Q4 leg, ISTM that the base voltage of Q3
will be nearly the same as that of Q2, which is driven from a 620
ohm / 22 ohm divider (plus various V_BEs). That makes the current
through the 220 ohm hard to estimate by eyeball. (The average
current is obviously going to be small, on account of that 100k
resistor.)
I'll have to introduce some notation to explain what's going on.
Isn is the speaker current supplied by the NPN Darlington.
Isp is the speaker current drawn in reverse direction by the PNP
Darlington.
Ven is NPN Darlington emitter voltage
Vep is PNP Darlington emitter voltage
IB is the common bias current supplied by the NPN emitter into the
PNP emitter sink.
R is the value of the current sense resistors, 0.33R 2W in the
schematic, no designators given.
Objective is to maintain IB constant, within reason, so as to
eliminate crossover distortion.
Total voltage across the 2 R's at any instant is the differential
Ven- Vep = R x ( Isn + IB + IB + Isp)
This rearranges to Ven- Vep = 2 x R x IB ( DC term) + R x ( Isn +
Isp) ( signal AC term)
Note R x Isn-p is Ven-p relative to Voutput AC.
Obviously a first requirement is to make d/dt ( Ven - Vep ) = 0,
making the DC term, and hence IB, a constant.
One simple way, and simplest is best, is to make Vep follow Ven
during the positive half output cycles, and Ven follow Vep during the negative half cycles.
And, viola- there you have your perfect bias with zero crossover.
IB is set by that current sink drawing a stiff current through the
27R in series with the LM311 (-) input.
Now you should be able to revisit your analysis of all those current
mirrors and their cascades to see how those followers are
implemented. Be sure to watch for the linearity of the LM311 with its
spec'd 200 V/mv gain.
(I sort of gather that it's pretty tweaky, due to that scribbled-in
10k to ground from the bases of Q3 and Q4.)
Did he say he built this?
I'm pretty sure the performance claimed was all SPICE derived, and
not actually measured.
On Friday, December 15, 2023 at 1:45:17 AM UTC-5, Phil Hobbs wrote:RL
On 2023-12-14 20:41, Fred Bloggs wrote:
On Monday, December 11, 2023 at 6:06:08 PM UTC-5, Phil Hobbs
wrote:
On 2023-12-11 12:09, Fred Bloggs wrote:
On Monday, December 11, 2023 at 10:50:04 AM UTC-5, legg
wrote:
On Mon, 11 Dec 2023 15:01:25 +0000, Clive Arthur
<cl...@nowaytoday.co.uk> wrote:
On 11/12/2023 14:05, legg wrote:
On Wed, 06 Dec 2023 23:22:26 GMT, Glen Walpert
<nos...@null.void> wrote:
<snip>
Some years ago Jim Thompson posted an audio
amplifier design which used current mirrors to
provide bias to the output transistors for the
express purpose of keeping crossover distortion low
over a large temperature range. He claimed it was the
bees knees, but a quick search failed to turn it up.
Perhaps someone else saved it or remembers the
thread?
Glen
From old LTSpice trash here;
http://ve3ute.ca/query/Half_Bridge_for_77_280Z_thompson.pdf
Nice arm waving there, 'Fred'. How exactly do you propose to doI'm not sure that we're looking at the same schematic. In theA lot of the bumph is dedicated only to biasing and it
Thanks!
Looks like the top output Darlington is AC coupled and
when the comparator detects a quiescent current through
the output resistors transitioning to less than some
value, it pumps the upper Darlington base voltage up a
bit, otherwise, the upper Darlington base voltage drifts
down.
Is that about right?
Not sure it would work in my application as my signal
isn't continuous - it spends some proportion of the time
idling at half supply. Still, I could probably arrange a
clock to force a comparator sample somehow.
Or maybe make the adjustment non-volatile (digipot?) and
clock it both up and down. The signal comes from a DAC,
so I do have access to timing signals.
would take some doing to get it to work over temperature
given those polarized cap sizes. Integrated darlingtons are
also best avoided. By 'wide range', the author was talking
standard industrial temperatures.
You'd also have to do some thin'in around the gain-setting
regime. Doubt this was a consideration in this drawing ( .
. . 'or' . . .), nor was 100KHz ( hence zobel network ).
I don't see quiescent conditions being an issue, but
start-up and shutdown could be surprising. Not sure that
was Thompson's strong point.
RL
That 100u base-to-base bias cap is probably needed to
dominant poll stabilize the LM311 amp more than anything
else.
one I have, "Half_Bridge_for_77_280Z_thompson.pdf", the LM311
is a weird sort of switching bias supervisor, not a linear amp.
It's running as a normal open-collector comparator, with its
output wire-ORed with the shutdown transistor Q7.
That's crazy. For one thing, audio speakers do not respond well
to discontinuities in the drive voltage. They do things like
snap, crackle and pop.
When it fires, or the shutdown line is high, it steals Q3's
base bias. That causes Darlington Q5/Q6 to turn on more, with a
TC of about 2 seconds. That reduces the quiescent bias.
In small-signal conditions, that'll just oscillate irregularly
and keep the class-A bias current of very roughly 60 mA. (*)
In large-signal conditions, the comparator will be pulling low
most of the time, which reduces the quiescent bias
progressively. (If the gain of the bias loop is high enough, it
may not drift that far, but I'd probably need to use SPICE to
find that out.)
When the shutdown pin is active, the class-A bias will
gradually go to 0, turning the output totem pole into a really
bad class B. (One gathers that the shutdown turns off the audio
input as well.)
One aspect that I don't understand well is Q3. With a 3k/100
ohm voltage divider in the Q4 leg, ISTM that the base voltage
of Q3 will be nearly the same as that of Q2, which is driven
from a 620 ohm / 22 ohm divider (plus various V_BEs). That
makes the current through the 220 ohm hard to estimate by
eyeball. (The average current is obviously going to be small,
on account of that 100k resistor.)
I'll have to introduce some notation to explain what's going on.
Isn is the speaker current supplied by the NPN Darlington.
Isp is the speaker current drawn in reverse direction by the PNP
Darlington.
Ven is NPN Darlington emitter voltage
Vep is PNP Darlington emitter voltage
IB is the common bias current supplied by the NPN emitter into
the PNP emitter sink.
R is the value of the current sense resistors, 0.33R 2W in the
schematic, no designators given.
Objective is to maintain IB constant, within reason, so as to
eliminate crossover distortion.
Total voltage across the 2 R's at any instant is the
differential Ven- Vep = R x ( Isn + IB + IB + Isp)
This rearranges to Ven- Vep = 2 x R x IB ( DC term) + R x ( Isn
+ Isp) ( signal AC term)
Note R x Isn-p is Ven-p relative to Voutput AC.
Obviously a first requirement is to make d/dt ( Ven - Vep ) = 0,
making the DC term, and hence IB, a constant.
One simple way, and simplest is best, is to make Vep follow Ven
during the positive half output cycles, and Ven follow Vep during
the negative half cycles.
And, viola- there you have your perfect bias with zero
crossover.
that?
The circuit idea can be modernized, making for a big improvement over
that nearly 50 year old instantiation.
Also it's the output current that makes the speaker move.
IB is set by that current sink drawing a stiff current throughGee, thanks. ;)
the 27R in series with the LM311 (-) input.
Now you should be able to revisit your analysis of all those
current mirrors and their cascades to see how those followers
are implemented. Be sure to watch for the linearity of the LM311
with its spec'd 200 V/mv gain.
(I sort of gather that it's pretty tweaky, due to that
scribbled-in 10k to ground from the bases of Q3 and Q4.)
Did he say he built this?
I'm pretty sure the performance claimed was all SPICE derived,
and not actually measured.
Not in 1977, it wasn't. The first public release of the original
Berkeley SPICE2 program was in 1972, and it was pretty
primitive--FORTRAN, punch cards, running on a CDC 6400. SPICE
didn't actually become useful until the early '80s.
Yeah, it was a piece of junk, and a mess, written by undergrad
students at UCB.
The big semi houses most certainly had their proprietary IC sims
running well before SPICE.
I'm sure Motorola had ample resources in this vein available at the
time.
On Monday, December 11, 2023 at 6:49:30 PM UTC, Glen Walpert wrote:
On Mon, 11 Dec 2023 15:01:25 +0000, Clive Arthur wrote:
On 11/12/2023 14:05, legg wrote:There was a lot of discussion of this circuit when it was posted, and Jim
On Wed, 06 Dec 2023 23:22:26 GMT, Glen Walpert <nos...@null.void>
wrote:
<snip>
Some years ago Jim Thompson posted an audio amplifier design which
used current mirrors to provide bias to the output transistors for the >>>>> express purpose of keeping crossover distortion low over a large
temperature range. He claimed it was the bees knees, but a quick
search failed to turn it up. Perhaps someone else saved it or
remembers the thread?
Glen
From old LTSpice trash here;
http://ve3ute.ca/query/Half_Bridge_for_77_280Z_thompson.pdf
RL
Thanks!
Looks like the top output Darlington is AC coupled and when the
comparator detects a quiescent current through the output resistors
transitioning to less than some value, it pumps the upper Darlington
base voltage up a bit, otherwise, the upper Darlington base voltage
drifts down.
Is that about right?
posted some models and simulations possibly still available on Phil's
archive. I don't have time to actually think about it right now, but here
are some post snips with comments and model links, sorry about the length: >>
------------
Here's half of the full H-bridge amplifiers that I built for my 1977
280Z... Image scanned in quarters and pieced together for easier
understanding...
<http://www.analog-innovations.com/SED/Half_Bridge_for_77_280Z.pdf>
--
If both output transistors are briefly off or very nearly off while the
output is increasing through crossover (zero or near zero current through >>> both emitter resistors), then the LM311 goes high due to the lag through >>> the RC on the negative input, delivering additional current to the
current mirror with illegible designations through D1, pulling current
from the 20uF 10V capacitor, increasing it's voltage thus increasing the >>> bias offset provided by Q5 and Q6 until there is enough bias voltage
difference to insure some small overlap in the on time of the output
transistors. Q1 and Q2 appear to keep the bias voltages centered between >>> the rails, and possibly Q8 pulls the negative input of the comparator
down enough to prevent noise from turning it on with no input?. (Not at
all sure about Q8, it might do more than that).
Am I close? Hints on Q8?
Regards,
Glen
You are virtually on the money!
Q8 is just a current mirror operating on R13 to establish the bias current >> at the zero crossing (your observation that corrections only occur while
passing thru the zero crossing are dead-on... except that the Q8 current
prevents both off).
...Jim Thompson
The Q5/Q6 Darlington is simply to knock down the base current so that a
long R/C time constant dominates.
...Jim Thompson
What's the SPICE quiescent bias? Back of the envelope, I get 75 or 80mA. >>>
How do you pick R15/R16/C4? Looks like it's to bootstrap the bias above
the 13.3V rail with a time constant longer than the roll-off of the
amplifier.
Best regards,
Spehro Pefhany
In a later life I might have used a diode. We improve our skill-set over
the years... at least some of us do... some just bloviate >:-}
...Jim Thompson
See...
As requested, entered into PSpice and simulated....
<http://www.analog-innovations.com/SED/My_1977_Z_Amp.pdf>
for the simulation (and a readable schematic).
Betwixt the "honey-do", I ran intermod distortion, comparing class-B to my >> class-A-B method, zip file now updated...
<http://www.analog-innovations.com/SED/JimThompsons_A-B-
Bias_Amplifier.zip>
...Jim Thompson
To go along with that schematic, here is the subcircuit that should work
in all modern flavors of Spice...
<http://www.analog-innovations.com/SED/My_1977_Z_Amp.zip>
To simulate my circuit in LTspice, open a text editor and type the
following...
* Jim Thompson's 1977 Z Amplifier *
** Analysis setup **
.tran 0 10m 0 100n .OPTIONS ITL1=1500 .OPTIONS ITL2=2000 .OPTIONS
ITL4=1000 .OPTIONS STEPGMIN .OP X1 IN OUT VCC 0 My_1977_Z_Amp VCC VCC 0
13.3V VIN IN 0 SIN 0 4 1K 0 0 0 .INC
"C:\InsertYourPathToCopyOf\My_1977_Z_Amp.sub"
*
.END
Save as whatever name rings your chime, say...
"JimThompson'sMarvelousAmplifier.cir" >:-}
Then open LTspice. On the Tools/Control Panel/Save Defaults section check
both Save Subcircuits... check-boxes.
Then Open "JimThompson'sMarvelousAmplifier.cir"
Then Run
View whatever node voltage or device current you like.
Irrespective of Larkin's stone throwing, it doesn't fail for several
reasons... one specifically because it was 1977. Can anyone guess what
that was?
Interestingly it takes LTspice _much_longer_ to run this circuit than it
does PSpice, particularly the bias point calculation is butt slow.
Note that you _do_not_ need to draw a schematic in LTspice (or any other
Spice, for that matter) to simulate someone else's circuit. Many of my
clients only have LTspice, so I just pass them a PDF schematic and a
netlist, and they can verify my work just fine.
...Jim Thompson
Latest version....Bias_Amplifier.zip>
<http://www.analog-innovations.com/SED/JimThompsons_A-B-
Turns out that my A-B bias is STUNNINGLY better than the conventional
diode-biased class-B... almost 30dB better on intermod distortion!
Intermod is what gives you those nasty atonal ear-piercing sounds when you >> play a Mozart wood-wind ensemble with French horn accompaniment.
After 36 years, revisiting my scheme, and fixing the bias droop, it's time >> for me to go back and roll my own sound system from scratch... like I did
up until my late 30's... then I got "too busy" ;-)
I'll toss the TL091 and put some discrete's in there... maybe even use my
TL431 diff-pair >:-}
...Jim Thompson
Here is a link to Mr Thompson's file (JimThompsons_A-B- Bias_Amplifier.zip) should anyone want to simulate the circuit.
https://1drv.ms/u/s!AkjNCaVyTfIag2w_3ym3b15HkMyP?e=PWxkbm
On Tue, 19 Dec 2023 21:53:41 -0500, Phil Hobbs <pcdhSpamMeSenseless@electrooptical.net> wrote:
On 2023-12-11 15:32, JM wrote:
On Monday, December 11, 2023 at 6:49:30?PM UTC, Glen Walpert wrote:
On Mon, 11 Dec 2023 15:01:25 +0000, Clive Arthur wrote:
On 11/12/2023 14:05, legg wrote:There was a lot of discussion of this circuit when it was posted, and Jim >>>> posted some models and simulations possibly still available on Phil's
On Wed, 06 Dec 2023 23:22:26 GMT, Glen Walpert <nos...@null.void>
wrote:
<snip>
Some years ago Jim Thompson posted an audio amplifier design which >>>>>>> used current mirrors to provide bias to the output transistors for the >>>>>>> express purpose of keeping crossover distortion low over a large >>>>>>> temperature range. He claimed it was the bees knees, but a quick >>>>>>> search failed to turn it up. Perhaps someone else saved it or
remembers the thread?
Glen
From old LTSpice trash here;
http://ve3ute.ca/query/Half_Bridge_for_77_280Z_thompson.pdf
RL
Thanks!
Looks like the top output Darlington is AC coupled and when the
comparator detects a quiescent current through the output resistors
transitioning to less than some value, it pumps the upper Darlington >>>>> base voltage up a bit, otherwise, the upper Darlington base voltage
drifts down.
Is that about right?
archive. I don't have time to actually think about it right now, but here >>>> are some post snips with comments and model links, sorry about the length: >>>>
------------
Here's half of the full H-bridge amplifiers that I built for my 1977
280Z... Image scanned in quarters and pieced together for easier
understanding...
<http://www.analog-innovations.com/SED/Half_Bridge_for_77_280Z.pdf>
--
If both output transistors are briefly off or very nearly off while the >>>>> output is increasing through crossover (zero or near zero current through >>>>> both emitter resistors), then the LM311 goes high due to the lag through >>>>> the RC on the negative input, delivering additional current to the
current mirror with illegible designations through D1, pulling current >>>> >from the 20uF 10V capacitor, increasing it's voltage thus increasing the >>>>> bias offset provided by Q5 and Q6 until there is enough bias voltage >>>>> difference to insure some small overlap in the on time of the output >>>>> transistors. Q1 and Q2 appear to keep the bias voltages centered between >>>>> the rails, and possibly Q8 pulls the negative input of the comparator >>>>> down enough to prevent noise from turning it on with no input?. (Not at >>>>> all sure about Q8, it might do more than that).
Am I close? Hints on Q8?
Regards,
Glen
You are virtually on the money!
Q8 is just a current mirror operating on R13 to establish the bias current >>>> at the zero crossing (your observation that corrections only occur while >>>> passing thru the zero crossing are dead-on... except that the Q8 current >>>> prevents both off).
...Jim Thompson
The Q5/Q6 Darlington is simply to knock down the base current so that a >>>> long R/C time constant dominates.
...Jim Thompson
What's the SPICE quiescent bias? Back of the envelope, I get 75 or 80mA. >>>>>
How do you pick R15/R16/C4? Looks like it's to bootstrap the bias above >>>>> the 13.3V rail with a time constant longer than the roll-off of the
amplifier.
Best regards,
Spehro Pefhany
In a later life I might have used a diode. We improve our skill-set over >>>> the years... at least some of us do... some just bloviate >:-}
...Jim Thompson
See...
As requested, entered into PSpice and simulated....
<http://www.analog-innovations.com/SED/My_1977_Z_Amp.pdf>
for the simulation (and a readable schematic).
Betwixt the "honey-do", I ran intermod distortion, comparing class-B to my >>>> class-A-B method, zip file now updated...
<http://www.analog-innovations.com/SED/JimThompsons_A-B-
Bias_Amplifier.zip>
...Jim Thompson
To go along with that schematic, here is the subcircuit that should work >>>> in all modern flavors of Spice...
<http://www.analog-innovations.com/SED/My_1977_Z_Amp.zip>
To simulate my circuit in LTspice, open a text editor and type the
following...
* Jim Thompson's 1977 Z Amplifier *
** Analysis setup **
.tran 0 10m 0 100n .OPTIONS ITL1=1500 .OPTIONS ITL2=2000 .OPTIONS
ITL4=1000 .OPTIONS STEPGMIN .OP X1 IN OUT VCC 0 My_1977_Z_Amp VCC VCC 0 >>>> 13.3V VIN IN 0 SIN 0 4 1K 0 0 0 .INC
"C:\InsertYourPathToCopyOf\My_1977_Z_Amp.sub"
*
.END
Save as whatever name rings your chime, say...
"JimThompson'sMarvelousAmplifier.cir" >:-}
Then open LTspice. On the Tools/Control Panel/Save Defaults section check >>>> both Save Subcircuits... check-boxes.
Then Open "JimThompson'sMarvelousAmplifier.cir"
Then Run
View whatever node voltage or device current you like.
Irrespective of Larkin's stone throwing, it doesn't fail for several
reasons... one specifically because it was 1977. Can anyone guess what >>>> that was?
Interestingly it takes LTspice _much_longer_ to run this circuit than it >>>> does PSpice, particularly the bias point calculation is butt slow.
Note that you _do_not_ need to draw a schematic in LTspice (or any other >>>> Spice, for that matter) to simulate someone else's circuit. Many of my >>>> clients only have LTspice, so I just pass them a PDF schematic and a
netlist, and they can verify my work just fine.
...Jim Thompson
Latest version....Bias_Amplifier.zip>
<http://www.analog-innovations.com/SED/JimThompsons_A-B-
Turns out that my A-B bias is STUNNINGLY better than the conventional
diode-biased class-B... almost 30dB better on intermod distortion!
Intermod is what gives you those nasty atonal ear-piercing sounds when you >>>> play a Mozart wood-wind ensemble with French horn accompaniment.
After 36 years, revisiting my scheme, and fixing the bias droop, it's time >>>> for me to go back and roll my own sound system from scratch... like I did >>>> up until my late 30's... then I got "too busy" ;-)
I'll toss the TL091 and put some discrete's in there... maybe even use my >>>> TL431 diff-pair >:-}
...Jim Thompson
Here is a link to Mr Thompson's file (JimThompsons_A-B- Bias_Amplifier.zip) should anyone want to simulate the circuit.
https://1drv.ms/u/s!AkjNCaVyTfIag2w_3ym3b15HkMyP?e=PWxkbm
I went do download this, but it wants a login.
Cheers
Phil Hobbs
I've emailed it to you. Links posted here will only be active for a
few days. If anyone else needs it try :-
https://1drv.ms/u/s!AkjNCaVyTfIahAKhXsApoGWJlEtu?e=ey7AEC
I did actually build a variation of this when it was originally
posted.
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